Tunable microwave devices with auto-adjusting matching circuit

ABSTRACT

An embodiment of the present disclosure provides a circuit including an antenna having a tunable component. The tunable component can be operable to receive a variable signal to cause the tunable component to change a reactance of the antenna. The tunable component can include a first conductor coupled to the antenna, a second conductor, and a tunable material positioned between the first conductor and the second conductor, where at least one of the first conductor or the second conductor, or both are adapted to receive the variable signal to cause the change in the reactance of the antenna. Additional embodiments are disclosed.

CROSS REFERENCED TO RELATED APPLICATIONS

This application is a continuation of application Ser. No. 13/745,624,entitled TUNABLE MICROWAVE DEVICES WITH AUTO-ADJUSTING MATCHING CIRCUIT,filed Jan. 18, 2013, which is a continuation of application Ser. No.12/952,395, entitled TUNABLE MICROWAVE DEVICES WITH AUTO-ADJUSTINGMATCHING CIRCUIT, filed Nov. 23, 2010, which is a continuation ofapplication Ser. No. 11/245,898, now U.S. Pat. No. 7,865,154, entitledTUNABLE MICROWAVE DEVICES WITH AUTO-ADJUSTING MATCHING CIRCUIT, filedOct. 8, 2005, which is a continuation in part of application Ser. No.10/938,898 entitled “TUNABLE MICROWAVE DEVICES WITH AUTO-ADJUSTINGMATCHING CIRCUIT”, filed Sep. 10, 2004, which is a continuation ofapplication Ser. No. 10/455,901, now U.S. Pat. No. 6,864,757, entitled“TUNABLE MICROWAVE DEVICES WITH AUTO-ADJUSTING MATCHING CIRCUIT”, filedJun. 6, 2003, which is a divisional of application Ser. No. 09/909,187,now U.S. Pat. No. 6,590,468, entitled “TUNABLE MICROWAVE DEVICES WITHAUTO-ADJUSTING MATCHING CIRCUIT”, filed Jul. 19, 2001, which claims thebenefit of U.S. provisional application No. 60/219,500, filed Jul. 20,2000. All sections of application Ser. Nos. 13/745,624, 12/952,395, andU.S. Pat. No. 7,865,154 are incorporated herein by reference in theirentirety.

BACKGROUND

Wireless communications is a rapidly growing segment of thecommunications industry, with the potential to provide high-speedhigh-quality information exchange between portable devices locatedanywhere in the world. Potential applications enabled by this technologyinclude, but are not limited to, multimedia Internet-enabled cellphones, smart homes and appliances, automated highway systems, videoteleconferencing and distance learning, and autonomous sensor networks,to name just a few. However, supporting these applications usingwireless techniques poses a significant technical challenge.

Presently, mobile phones employing the Global System for MobileCommunication (‘IGSM”) standard operate in multiple frequency bands.Mobile phones may be capable of three or four frequency bands, therebyallowing the mobile phone to be used with a variety of serviceproviders. However, the speed and quality of GSM will not meet therequirements of the large data transmission of the future.

GPRS (General Packet Radio Services) is a packet-based wireless thatpromises data rates from 56 up to 114 Kbps and continuous connection tothe Internet for mobile phone and computer users. GPRS is based on“regular” GSM (with the same modulation) and will complement existingservices such circuit-switched cellular phone connections such as SMS orcell broadcast. Voice over IP over GPRS has also been explored.

Similarly, Enhanced Data rate for Global Evolution (EDGE) is a radiobased high-speed mobile data standard. It allows data transmissionspeeds of 384 kbps to be achieved when all eight timeslots are used.This means a maximum bit rate of 48 kbps per timeslot. Even higherspeeds and bandwidths are now available as WCDMA (Wideband Code DivisionMultiple Access) is implemented.

As handsets move to meet increased bandwidth needs, the requirements ofcomponents are more astringent. Battery life has to be maximized,reception clarity in a multitude of environments has to be improved andat the same time the customers require a significant reduction in size.These requirements burden the Radio Frequency (RF) front end modules(FEM) to be capable of modulating efficiently and dynamically betweenthe leading global wireless broadband standards in a single platformsolution.

I. Impedance Match

In order to maximize battery life, energy transfer received by RF(inputs and outputs) must be perfectly matched to the impedance of theantennas. This impedance is typically assumed to be 50 ohms. For nomismatch losses to occur, the impedance of the radio must be the“conjugate match” of the impedance of the antenna. That is, theimpedances must have the exact same value of real impedance (resistivepart) and opposite but equal value of imaginary impedances (reactivepart). For example, if the radio impedance is 40+j6, the antennaimpedance must be 40−j6 in order to have no power lost due to impedancemismatch.

However, this 50 ohm design target is never perfectly met and also anantenna's impedance is never consistently 50 ohms over all conditionsand frequencies of operation. The greater the difference in impedance,the less power reaches the antenna and radiates. Instead, a portion ofthis power (proportional to the impedance mismatch) is reflected backtowards the radio during transmission and never gets radiated out of theantenna as intended. Given the reciprocal nature of most radios withantennas, this mismatch reduces the amount of power received by theradio from the antenna during reception. Because the mobile phone isoperated in various environments, the antenna is “detuned”, resulting inimpedance at its I/O port which can vary wildly. Similarly, unit-to-unitvariations in performance can be seen in power amplifiers, filters,switches and antennas when produced in high volume. Regardless of whythe two impedances may not be matched, the greater the difference, thegreater the power level reflected and the lower the power transferred(transmitted or received), thus lowering the overall efficiency of theRF system (antenna+radio). Also, antennas often compromise bandwidth inexchange for greater gain (efficiency) or reduced size. This results ingreater variation of its impedance over the frequency of operation.

II The Voltage Standing Wave Ratio (VSWR)

The voltage standing wave ratio is a measure of how well a load isimpedance-matched to a source. The value of VSWR is always expressed asa ratio with 1 in the denominator (2:1, 3:1, 10:1, etc.). It is a scalarmeasurement, so although they reflect waves 15 oppositely, a shortcircuit and an open circuit have the same VSWR value (infinity: 1). Aperfect impedance match corresponds to a VSWR 1:1, but in practice itwill never be achieved. The reflection coefficient can be read from aSmith chart. A reflection coefficient magnitude of zero is a perfectmatch; a value of one is perfect reflection. The return loss of a loadis merely the magnitude of the reflection coefficient expressed indecibels.

III Existing Methodologies

A. Fixed Impedance Networks.

With a fixed matching network, an amplifier which had been tuned forpeak efficiency at the maximum output power level will show severelydegraded efficiency as the gain is lowered and the output power drops.Standard matching networks present fixed impedance transformations at agiven frequency. This limits the efficient operating range of anamplifier to 5 a narrow range of frequencies and a small range of powerand bias settings.

B. Digital Matching Networks

Digital impedance matching using hybrid transformers, discrete OpAmpcircuits, integrated analog/digital implementations are used tosynthesize the matching impedance. Digital implementations of impedancematching networks are problematic due to the delays through the digitalprocessors. These delays cause a phase difference between the incidentand synthesized signals which results in an amplitude difference thatworsens the return loss. This problem is generally alleviated by using afaster, but more costly and higher power consuming digital converter andprocessor.

C. Adaptive or Variable Matching Networks

It is possible to provide an adaptive antenna matching network whichcompensates for these variations. Several advantages offered by tunablematching networks, including optimization of amplifier match at varyingfrequency and against variable load VSWR. As the output power is reducedin a variable gain amplifier, the load impedance which allows theamplifier to operate with optimum efficiency changes. A system monitorsthe signal strength of an incoming signal, to provide indication ofsignal strength. This signal strength level is also used to adapt anadaptive network arranged to compensate for variations in antennaimpedance. A problem with the above method is that adaptations of theantenna matching network may be effected in response to reductions insignal strength, when said reductions occur due to reasons other thanantenna mismatch; for example, when the variations are due to mismatchin several mobile standards such as switching from GSM mode to EDGEmode. Therefore, adaptations do not improve transmission characteristicsand at worst may make the transmission characteristics worse.

In variable impedance matching networks, the impedance can be adjustedto allow optimum efficiency at both low and high power settings. Outputmatching networks consisting of lumped capacitors (standard ceramic ortantalum capacitors), varactors, inductors, transmission lines, ortransformers, suffer because the loss associated with the individualcomponents. At the RF frequencies the more components used in thenetwork, the aggregate of the individual component loss will worsen thetotal loss of the matching network.

D. Tunable Capacitors

High performance tunable capacitors are frequently used to tune variableimpedance matching networks for RF communication applications. Typicalcapacitance values for these applications range from 1 pF to 5 pF with atuning range of approximately 20% for Voltage Controlled Oscillators(VCOs) and 100% for antennas and filters. High quality factors 20 (Qs)over 50 are critical for these high performance tuning applications. Alarge voltage swing capability is highly desirable for implementingtransmitter building blocks, where an RF voltage amplitude can reach 20V peak to peak for an output power of 1 W.

Conventional tunable capacitors, commonly referred to as varactordiodes, are implemented using silicon PN-junctions and Metal OxideSemiconductor (MOS) capacitor structures. These devices suffer from alimited qualify factor and tuning range, and exhibit a low voltage swingcapability limited by the forward biasing of PN junctions or silicondioxide breakdown, thus inadequate for high performance RF applications.The most commonly used varactor is a semiconductor diode varactor.However, semiconductor diode varactors suffer from low Q factors(especially at high frequencies), low power handling, and highintermodulation distortion. Common varactors used today are silicon andgallium arsenide based diodes.

Micromachined tunable capacitors have been proposed to improve variousaspects of the device performance. Aluminum micromechanical tunablecapacitors have been demonstrated by vertically moving a suspended platetowards the substrate with an electrostatic force. The device achieves ahigh Q due to low resistive loss and a moderate tuning range adequatefor low phase noise RF VCOs.

Copper-based micromachined tunable capacitors, relying on laterallyactuating a dielectric slab between two sets of copper electrodes, havebeen developed to obtain an improved quality factor but suffer from asmall tuning range due to the dielectric layer warpage caused by thefilm strain gradient.

Silicon-based micromachined tunable capacitors have also beeninvestigated, including vertically actuating a movable plate between twofixed electrodes and laterally varying the overlap area between two setsof comb drive fingers to achieve a large tuning range around 100% with a5 V tuning voltage. However, due to the excessive silicon resistiveloss, the devices exhibit low quality factors at high frequencies.

Recently, micromachined tunable capacitors operating in dielectric fluidhave also been demonstrated to achieve an increased capacitance densitywith improved tuning performance. However, besides the low qualityfactor caused by the large silicon resistive loss, complex fluidicpackaging is required not suitable for mobile applications.

Ferroelectric materials can also be used to create matching networks forfixed as well as variable impedance matching applications. Thedielectric constant of the ferroelectric material can be varied by a DCvoltage applied across the ferroelectric. Because the dielectricconstant of the ferroelectric material changes over time due to severalfactors including temperature, humidity, aging of the material andhysteresis, it has to be well known before and calibrated during thefabrication process. The degree to which temperature, humidity, aging,hysteresis and the like cause the value of the dielectric constant tovary over time depends on the particular material. Ferroelectricmaterials are available which demonstrate minimal dielectric constantvariations as a result of these factors. However, such materials havethe undesirable properties of low phase change with voltage which is themain parameter of interest between the conductor line and the groundplane.

Recent advances in tunable ferroelectric materials have allowed forrelatively low capacitance varactors that can operate at temperaturesabove those necessary for superconduction and at bias voltages less thanthose required for existing planar varactor structures, whilemaintaining high tunability and high Q factors. Even though thesematerials work in their paraelectric phase above the Curie temperature,they are conveniently called “ferroelectric” because they exhibitspontaneous polarization at temperatures below the Curie temperature.Tunable ferroelectric materials including barium-strontium titanateBaxSrl-x Ti03 CBST) or BST composites have been the subject of severalpatents. Dielectric materials including BST are disclosed by Sengupta,et al. in U.S. Pat. No. 5,312,790; U.S. Pat. No. 5,427,988; U.S. Pat.No. 5,486,491; U.S. Pat. No. 5,846,893; U.S. Pat. No. 5,635,434; U.S.Pat. No. 5,830,591; U.S. Pat. No. 5,766,697; U.S. Pat. No. 5,693,429;U.S. Pat. No. 6,074,971; U.S. Pat. No. 6,801,104 B2 and U.S. Pat. No.5,635,433. These patents are hereby incorporated by reference. Thepermittivity (more commonly called dielectric constant) of thesematerials can be varied by varying the strength of an electric field towhich the materials are subjected. These materials allow for thin-filmferroelectric composites of low overall dielectric constant that takesadvantage of the high tunability and at the same time having highdielectric constants.

Embedded mobile phone antennas have significant performance issues suchas dramatic drop in efficiencies when operating conditions change fromfree space (in a hands-free cradle) to in-situ (against head/hand), withtypical efficiencies going from 70% to 25% for the high bands and from50% dropping down to 10% for the low bands. Also antenna bandwidthsoften compromises band-edge performance, especially for antennas havingaggressive size reduction, the laws of physics limits gain bandwidthproduct possible for a given antenna volume.

For the foregoing reasons, there is a need in the RF industry to developa self-contained dynamic, tunable (variable) matching antenna networkfor improved efficiency in the “de-tuned” state of components, withminimal mismatch loss in multi-mode operation. To match dynamicallymultiple power levels, temperature changes, aging of components in RFcircuits. A matching antenna network compliant to many different highefficiency power amplifier front end modules (FEMs), for multipleprotocols in a multi-mode/multi band/multi-frequency devices.

SUMMARY OF THE INVENTION

An embodiment of the present invention provides an apparatus, comprisingan input port and a dynamic impedance matching network capable ofdetermining a mismatch at the input port and dynamically changing the RFmatch by using at least one matching element that includes at least onevoltage tunable dielectric capacitor. The matching network may be a“Pi”, a “T”, or “ladder” type network and the apparatus may furthercomprise at least one directional coupler capable of signal collectionby sampling a portion of an incident signal, a reflected signal or both.In an embodiment of the present invention, the apparatus may alsoinclude a control and power control & logic unit (PC LU) to convertinput analog signals into digital signals and sensing VSWR phase andmagnitude and processing the digital signals using an algorithm to giveit a voltage value and wherein the voltage values may be compared tovalues coming from the coupler and once compared and matched, the valuesmay be passed to a Hi Voltage Application Specific Integrated Circuit(HV ASIC) to transfer and distribute compensatory voltages to thematching network elements.

In an embodiment of the present invention the sampling may beaccomplished by at least one low impedance capacitor sampling elementand the sampling element may be an interdigital capacitor ormetal-insulator-metal (MIM) capacitor.

In an embodiment of the present invention, the aforementioned “ladder”type network may comprise four inductors in series and may include threegrounded matching capacitors in parallel connected to the inductors.

In an embodiment of the present invention, the dynamic impedancematching network may be mounted onto a low-cost thermally conductivedielectric substrate with components surface mounted on one side, thenover-molded as a Multi-Chip-Module (MCM), which may be itself surfacemountable on to a printed wiring board of an RF devise. Further, thedynamic impedance matching network system may be positioned between aPower Amplifier (P A) unit and a frequency filter unit in the apparatusand the apparatus may be a mobile phone and further comprises anexternal digital signal processor to convert input analog signals intodigital signals and sensing VSWR phase and magnitude and processing thedigital signals using an algorithm to give it a voltage value andwherein the voltage values are compared to values coming from thecoupler and once compared and matched, compensatory voltages aretransferred and distributed to the matching network elements.

In an embodiment of the present invention, the apparatus may be a mobilephone and the dynamic impedance matching network system may bepositioned between a frequency filter unit and a switch unit in themobile phone or the dynamic impedance matching network may be a singlestand-alone component that may be placed in the transmit chain of amobile phone radio. Also, the apparatus may be a mobile phone and thedynamic impedance matching network may be self-contained therebyrequiring only a fixed DC bias 3V from the mobile phone. Thisself-contained dynamic impedance matching network system enablesimpedance matching and therefore operation in a GSM, EDGE, CDMA or WCDMAor multi-mode mobile phone, and at least capable of a single bandoperation that includes 800, 900, 1800, 1900 MHz bands and the 2.1 GHzband.

In another embodiment of the present invention is provided a method ofmatching the impedance of an RF signal received by an input port of adevice by using a dynamic impedance matching network capable ofdetermining a mismatch at the input port and dynamically changing the RFmatch by using at least one matching element that includes at least onevoltage tunable dielectric capacitor.

In yet another embodiment of the present invention is provided anauto-adjusting matching circuit, comprising a first and second couplerto sample an incoming signal and an outgoing signal, a matching networkto receive the incoming signal and including a plurality of matchingnetwork elements comprising voltage tunable dielectric capacitors,wherein the matching network is capable of automatically adjusting animpedance of the incoming signal, and a power control & logic unit toproduce a voltage to output ports from a High Voltage ApplicationSpecific Integrated Circuit to the plurality of matching networkelements.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention is described with reference to the accompanyingdrawings. In the drawings, like reference numbers indicate identical orfunctionally similar elements. Additionally, the left-most digit(s) of areference number identifies the drawing in which the reference numberfirst appears.

FIG. 1 illustrates a variable matching capacitor that may be used in anembodiment of the present invention;

FIG. 2 provides a block representation of on embodiment of the presentinvention;

FIG. 3 is a schematical representation of one embodiment of the presentinvention illustrating the use of a “Pi” network as the means for thematching network;

FIG. 4 is a schematical representation of one embodiment of the presentinvention, depicting the use of a “PP’ network as the means for thematching network;

FIG. 5 is a schematical representation of one embodiment of the presentinvention depicting the use of a “T” network as the means for thematching network;

FIG. 6 is a schematical representation of one embodiment of the presentinvention depicting the use of a “T” network as the means for thematching network;

FIG. 7 is a schematical representation of a “Ladder” network comprisingthree “T” networks for use as the means for the matching network or asreplacement for the matching network elements of one embodiment of thepresent invention;

FIG. 8 is a schematical representation of a network for use as the meansfor the matching network or as replacement for the matching networkelements of one embodiment of the present invention;

FIG. 9 is a schematical representation of a network for the use as themeans for the matching network or as replacement for the matchingnetwork elements with an inductor in parallel to a matching capacitor inan embodiment of the present invention;

FIG. 10 is a schematical representation of a variable matching capacitorcapable of being used as the means for the matching network elements andas replacement for any of the matching capacitors of one embodiment ofthe present invention;

FIG. 11 is a graphical block representation of the invention as it isused as part of a multi-band Radio Frequency (RF) system in oneembodiment of the present invention;

FIG. 12 provides a table of data collected from measurements of a RFcellular phone placed on a first test environment; and

FIG. 13 provides a table of data collected from measurements of a RFcellular phone placed on a second test environment.

DETAILED DESCRIPTION

In the following detailed description, numerous specific details are setforth in order to provide a thorough understanding of the invention.However, it will be understood by those skilled in the art that thepresent invention may be practiced without these specific details. Inother instances, well-known methods, procedures, components and circuitshave not been described in detail so as not to obscure the presentinvention.

An impedance matching network or circuit, may be a combination ofreactive elements connected between a circuit and at least one load thattransform the load impedance into another impedance value to achieveimproved performance such as maximum power transfer, reducedreflections, or optimum load performance. An impedance matching networkmay be made up of a combination of lumped elements, (resistors,capacitors, and inductors), or distributed elements (transmission linesof varying characteristic impedance and length). Similarly, an impedancematching network, transforms the circuit impedance from one value toanother. A varactor of one embodiment of the present invention that maybe used in an impedance matching network is illustrated in FIG. 1, withFIG. 2 depicting a block diagram showing a matching network 10constructed in accordance with an embodiment of the present inventioncoupled to a tunable microwave device 232. The tunable device 232 couldbe one of many devices which have varying input/output characteristicimpedances such as tunable phase shifters, delay lines, filters, etc. Inthe arrangement shown in FIG. 2, the adjustable external DC voltagesource is used to supply bias voltage to the matching network 10 and thetunable microwave device 232 in tandem. As the voltage supplied by theexternal DC voltage source changes, the characteristic input/outputimpedance of the tunable dielectric device will also change. At the sametime the impedance characteristics of the matching network will changeto maximize power transfer from/to the microwave source/load 234 to/fromthe tunable microwave device 232. Alternatively, the tunable microwavedevice 232 and the matching network 10 can be controlled by twodifferent external DV voltage sources 34.

It may be determined, based on required source and load reflectioncoefficients, the ideal matching network required for a particularcircuit. Depending on the center frequency, the proper individual valuesof the lumped elements or lengths of the transmission lines will beapplied. In general, there are several different matching circuitsavailable. Based on insertion loss, simplicity, and reproducibility ofeach matching element, the best selection can be made. Some of thepossible networks typically used are described but not limited to FIG.7, FIG. 8 and FIG. 9. For instance, many matching circuits may use shuntcapacitors (parallel plate) or shunt inductors (spiral inductors), andappreciable voltage exists across the shunted component, their mainreason is to provide a fixed voltage increase or to improve the powerfactor in the circuit. For example, on the input to a low noisetransistor, the impedance of an incoming 75 ohm transmission line wouldbe transformed by the input matching network to the impedance Zopt,required to achieve the minimum noise figure of the transistor. TheSmith chart is a tool commonly used by microwave engineers to aid withimpedance matching.

Turning now to FIG. 3 is illustrated a dynamic impedance matchingnetwork system 100 that is composed of a “Pi” network as the means forthe matching network 10. The “Pi” type network comprises matchingnetwork elements 10 a, 10 b, 10 c in which in an embodiment of thepresent invention may be of the BST type variable matching capacitors313 of FIG. 10.

Throughout the description of the present invention, BST may or has beenused as a tunable dielectric material that may be used in a tunabledielectric capacitor of the present invention. However, the assignee ofthe present invention, Paratek Microwave, Inc. has developed andcontinues to develop tunable dielectric materials that may be utilizedin embodiments of the present invention and thus the present inventionis not limited to using BST material. This family of tunable dielectricmaterials may be referred to as Parascan®.

The term Parascan® as used herein is a trademarked term indicating atunable dielectric material developed by the assignee of the presentinvention. Parascan® tunable dielectric materials have been described inseveral patents. Barium strontium titanate (BaTiO3-SrTiO3), alsoreferred to as BSTO, is used for its high dielectric constant(200-6,000) and large change in dielectric constant with applied voltage(25-75 percent with a field of 2 Volts/micron). Tunable dielectricmaterials including barium strontium titanate are disclosed in U.S. Pat.No. 5,312,790 to Sengupta, et al. entitled “Ceramic FerroelectricMaterial”; U.S. Pat. No. 5,427,988 by Sengupta, et al. entitled “CeramicFerroelectric Composite Material-BSTO-MgO”; U.S. Pat. No. 5,486,491 toSengupta, et al. entitled “Ceramic Ferroelectric CompositeMaterial—BSTO-Zr02”; U.S. Pat. No. 5,635,434 by Sengupta, et al.entitled “Ceramic Ferroelectric Composite Material-BSTO-Magnesium BasedCompound”; U.S. Pat. No. 5,830,591 by Sengupta, et al. entitled“Multilayered Ferroelectric Composite Waveguides”; U.S. Pat. No.5,846,893 by Sengupta, et al. entitled “Thin Film FerroelectricComposites and Method of Making”; U.S. Pat. No. 5,766,697 by Sengupta,et al. entitled “Method of Making Thin Film Composites”; U.S. Pat. No.5,693,429 by Sengupta, et al. entitled “Electronically Graded MultilayerFerroelectric Composites”; U.S. Pat. No. 5,635,433 by Sengupta entitled“Ceramic Ferroelectric Composite Material BSTO-ZnO”; U.S. Pat. No.6,074,971 by Chiu et al. entitled “Ceramic Ferroelectric CompositeMaterials with Enhanced Electronic Properties BSTO Mg BasedCompound-Rare Earth Oxide”. These patents are incorporated herein byreference. The materials shown in these patents, especially BSTO-MgOcomposites, show low dielectric loss and high tunability. Tunability isdefined as the fractional change in the dielectric constant with appliedvoltage.

Barium strontium titanate of the formula BaxSrl-xTiO3 is a preferredelectronically tunable dielectric material due to its favorable tuningcharacteristics, low Curie temperatures and low microwave lossproperties. In the formula BaxSrl-xTiO3, x can be any value from 0 to 1,preferably from about 0.15 to about 0.6. More preferably, x is from 0.3to 0.6.

Other electronically tunable dielectric materials may be used partiallyor entirely in place of barium strontium titanate. An example isBaxCal-xTiO3, where x is in a range from about 0.2 to about 0.8,preferably from about 0.4 to about 0.6. Additional electronicallytunable ferroelectrics include PbxZrl-xTiO3 (PZT) where x ranges fromabout 0.0 to about 1.0, PbxZrlxSrTiO3 where x ranges from about 0.05 toabout 0.4, KTaxNbl-xO3 where x ranges from about 0.0 to about 1.0, leadlanthanum zirconium titanate (PLZT), PbTiO3, BaCaZrTiO3, NaNO3, KNbO3,LiNbO3, LiTaO3, PbNb2O6, PbTa2O6, KSr(NbO3) and NaBa2(NbO3)5KH2PO4, andmixtures and compositions thereof. Also, these materials can be combinedwith low loss dielectric materials, such as magnesium oxide (MgO),aluminum oxide (AI2O3), and zirconium oxide (ZrO2), and/or withadditional doping elements, such as manganese (MN), iron (Fe), andtungsten (W), or with other alkali earth metal oxides (Le. calciumoxide, etc.), transition metal oxides, silicates, niobates, tantalates,aluminates, zirconnates, and titanates to further reduce the dielectricloss.

In addition, the following U.S. Patent Applications, assigned to theassignee of this application, disclose additional examples of tunabledielectric materials: U.S. application Ser. No. 09/594,837 filed Jun.15, 2000, entitled “Electronically Tunable Ceramic Materials” IncludingTunable Dielectric and Metal Silicate Phases”; U.S. application Ser. No.09/768,690 filed Jan. 24, 2001, entitled “Electronically Tunable,Low-Loss Ceramic Materials Including a Tunable Dielectric Phase andMultiple Metal Oxide Phases”; U.S. application Ser. No. 09/882,605 filedJun. 15, 2001, entitled “Electronically Tunable Dielectric CompositeThick Films And Methods Of Making Same”; U.S. application Ser. No.09/834,327 filed Apr. 13, 2001, entitled “Strain-Relieved TunableDielectric Thin Films”; and U.S. Provisional Application Ser. No.60/295,046 filed Jun. 1,2001 entitled “Tunable Dielectric CompositionsIncluding Low Loss Glass Frits”. These patent applications areincorporated herein by reference.

The tunable dielectric materials can also be combined with one or morenontunable dielectric materials. The non-tunable phase(s) may includeMgO, MgA12O4, MgTiO3, Mg2SiO4, CaSiO3, MgSrZrTiO6, CaTiO3, A12O3, SiO2and/or other metal silicates such as BaSiO3 and SrSiO3. The non-tunabledielectric phases may be any combination of the above, e.g., MgOcombined with MgTiO3, MgO combined with MgSrZrTiO6, MgO combined withMg2SiO4, MgO combined with Mg2SiO4, Mg2SiO4 combined with CaTiO3 and thelike.

Additional minor additives in amounts of from about 0.1 to about 5weight percent can be added to the composites to additionally improvethe electronic properties of the films. These minor additives includeoxides such as zirconnates, tannates, rare earths, niobates andtantalates. For example, the minor additives may include CaZrO3, BaZrO3,SrZrO3, BaSnO3, CaSnO3, MgSnO3, Bi2O3/2SnO2, Nd2O3, Pr7O11, Yb2O3,HO2O3, La2O3, MgNb2O6, SrNb2O6, BaNb2O6, MgTa2O6, BaTa2O6 and Ta2O3.

Thick films of tunable dielectric composites may comprise Ba1-xSrxTiO3,where x is from 0.3 to 0.7 in combination with at least one non-tunabledielectric phase selected from MgO, MgTiO3, MgZrO3, MgSrZrTiO6, Mg2SiO4,CaSiO3, MgA12O4, CaTiO3, A12O3, SiO2, BaSiO3 and SrSiO3. Thesecompositions can be BSTO and one of these components, or two or more ofthese components in quantities from 0.25 weight percent to 80 weightpercent with BSTO weight ratios of 99.75 weight percent to 20 weightpercent.

The electronically tunable materials may also include at least one metalsilicate phase. The metal silicates may include metals from Group 2A ofthe Periodic Table, i.e., Be, Mg, Ca, Sr, Ba and Ra, preferably Mg, Ca,Sr and Ba. Preferred metal silicates include Mg2SiO4, CaSiO3, BaSiO3 andSrSiO3. In addition to Group 2A metals, the present metal silicates mayinclude metals from Group 1A, i.e., Li, Na, K, Rb, Cs and Fr, preferablyLi, Na and K. For example, such metal silicates may include sodiumsilicates such as Na2SiO3 and NaSiO3-5H2O, and lithium-containingsilicates such as LiAISiO4, Li2SiO3 and Li4SiO4. Metals from Groups 3A,4A and some transition metals of the Periodic Table may also be suitableconstituents of the metal silicate phase. Additional metal silicates mayinclude A12Si2O7, ZrSiO4, Ka1Si3O8, NaAISi3O8, CaA12Si2O8, CaMgSi2O6,BaTiSi3O9 and Zn2SiO4. The above tunable materials can be tuned at roomtemperature by controlling an electric field that is applied across thematerials.

In addition to the electronically tunable dielectric phase, theelectronically tunable materials can include at least two additionalmetal oxide phases. The additional metal oxides may include metals fromGroup 2A of the Periodic Table, i.e., Mg, Ca, Sr, Ba, Be and Ra,preferably Mg, Ca, Sr and Ba. The additional metal oxides may alsoinclude metals from Group 1A, i.e., Li, Na, K, Rb, Cs and Fr, preferablyLi, Na and K. Metals from other Groups of the Periodic Table may also besuitable constituents of the metal oxide phases. For example, refractorymetals such as Ti, V, Cr, Mn, Zr, Nb, Mo, Hf, Ta and W may be used.Furthermore, metals such as AI, Si, Sn, Pb and Bi may be used. Inaddition, the metal oxide phases may comprise rare earth metals such asSc, Y, La, Ce, Pr, Nd and the like.

The additional metal oxides may include, for example, zirconnates,silicates, titanates, aluminates, stannates, niobates, tantalates andrare earth oxides. Preferred additional metal oxides include Mg2SiO4,MgO, CaTiO3, MgZrSrTiO6, MgTiO3, MgA12O4, WO3, SnTiO4, ZrTiO4, CaSiO3,CaSnO3, CaWO4, CaZrO3, MgTa2O6, MgZrO3, MnO2, PbO, Bi2O3 and La2O3.Particularly preferred additional metal oxides include Mg2SiO4, MgO,CaTiO3, MgZrSrTiO6, MgTiO3, MgA12O4, MgTa2O6 and MgZrO3.

The additional metal oxide phases are typically present in total amountsof from about 1 to about 80 weight percent of the material, preferablyfrom about 3 to about 65 weight percent, and more preferably from about5 to about 60 weight percent. In one preferred embodiment, theadditional metal oxides comprise from about 10 to about 50 total weightpercent of the material. The individual amount of each additional metaloxide may be adjusted to provide the desired properties. Where twoadditional metal oxides are used, their weight ratios may vary, forexample, from about 1:100 to about 100:1, typically from about 1:10 toabout 10:1 or from about 1:5 to about 5:1. Although metal oxides intotal amounts of from 1 to 80 weight percent are typically used, smalleradditive amounts of from 0.01 to 1 weight percent may be used for someapplications.

The additional metal oxide phases can include at least two Mg-containingcompounds. In addition to the multiple Mg-containing compounds, thematerial may optionally include Mg-free compounds, for example, oxidesof metals selected from Si, Ca, Zr, Ti, Al and/or rare earths. Thematching network elements 10 a, 10 c may be connected to matchingnetwork element 10 b by means of line 106 and line 107. These lumpedcomponents in the matching network 10 are ideally suited for integratedimpedance matching at low GHz frequencies. This type of matching network10 may be further preferred because it may comprises passive componentssuch as inductors and metal-BST or dielectric-metal capacitors with highquality factors not generating noise or loss as with resistive networks.It is understood that the present invention is not limited to theaforementioned passive components.

The signal collecting means used in FIG. 3 may be directional couplers101 and 102. These may be passive 3 or 4 port devices used to sample aportion of the forward (incident) signal or the reverse (reflected)signal, or both (dual directional coupler) in a RF, microwave circuit.Other types of couplers may be used as sampling means, such as branchline couplers, or a simple transmission line coupler and the presentinvention is not limited to any particular type of coupler. In anembodiment of the present invention, a transmission line coupler may becomposed of two transmission lines that allow signals to be coupled ortransferred in part from one line to the other.

The purpose of the coupler 102 may be to sample the incoming(mismatched) signals in one direction by means of line 110 and onreverse direction in line 109. In order to maximize coupling efficiencyand to increase coupling field, two coupling ports may be used 109 and110. There is transfer the RF energy from signal 104 to line 106 to thecoupling means of ports 110 and 109 into the control and power control &logic unit (PC LU) 116 which converts the input analog signals 104 intodigital signals, senses VSWR phase and magnitude, processes the digitalsignals using an algorithm and gives it a voltage value. These valuesmay be compared to values coming from coupler 101 through coupling ports105 and 108 into the logic and power control & logic unit (PC LU) 116.Once compared and matched, these values may later be fed through aconnection means 118 into a Hi Voltage Application Specific IntegratedCircuit (HV ASIC) 117 by which further transfers and distributes thecompensatory voltages to the matching network elements 10 a, 10 b and 10c.

In an antenna, the impedance is the ratio of the applied or inducedvoltage to the current flowing into or out of the antenna input. Moregenerally, it is defined as the ratio of the electric field to themagnetic field. The purpose of using variable matching capacitors 313 asmatching network elements 10 a, 10 b and 10 c is to compensate themismatched impedances in the RF circuit by means of dynamicallyincreasing or decreasing the voltage across said BST capacitors. Thecapacitance may be used to “tune” the RF circuit with an externallyapplied variable voltage 34. The algorithms in the power control & logicunit (PCLU) 116 may be based on dynamic equations (independentequations) for the purpose of control. Voltage compensation operationsapplied to the matching network elements 10 a, 10 b, and 10 c, may beemployed in a closed loop control system. This means will counteractdynamic lags or to modify the impedance between measured voltage ofcoupler 101 and coupler 102. Variables in the power control & logic unit(PCLU) 116 produces a prompt stable response voltage to output ports111, 112, 113 from the HV ASIC 117. The compensation action may beconditioned by speed of the power control & logic unit (PCLU) 116 andthe frequency of the incoming signals 104 and 103. In an embodiment ofthe present invention the variable capacitance of matching networkelements 10 a, 10 b and 10 c may allow for a simple low loss means todynamically compensate any RF circuit in an autonomous system loop.

According to another embodiment of the present invention as illustratedin FIG. 4 is a dynamic impedance matching network system 200 that iscomposed of a “Pi” network as the means for the matching network 10. The“Pi” type network may comprise matching network elements 10 a, 10 b, 10c in which in an embodiment of the present invention (and not limited inthis respect) to BST type variable matching capacitors 313 (as describedin FIG.1). The matching network elements 10 a, 10 c may be connected tomatching network element 10 b by means of line 106 and line 107. Theselumped components in matching network 10 are ideally suited forintegrated impedance matching at low GHz frequencies. This type ofmatching network 10in an embodiment of the present invention maycomprise passive components such as inductors and metal-BST ordielectric-metal capacitors with high quality factors not generatingnoise or loss as with resistive networks.

The signal sampler used in FIG. 4 may comprise sampling capacitors 204and 203 and may be advantageously used as low impedance capacitorsampling elements 201, 202. These may be passive devices used to samplethe voltage drop of a portion of the forward (incident) signal or thereverse (reflected) signal, or both in a RF, microwave circuit at anytwo points in the transmission line. The sampling capacitors 204 and 203may be used to estimate coupling between two circuit points to make surea minimum of coupling is obtained. Capacitance is dependent on conductorgeometry, conductor spatial relationships, and the material propertiessurrounding the conductors. Capacitors are usually constructed as twometal surfaces separated by a nonconducting material. Printed capacitorsform very convenient and inexpensive small capacitance values becausethey are printed directly on the printed circuit board (PCB) orsubstrate. For most applications, the higher the unloaded Q the betterthe capacitor. Gap capacitors are best used for very weak coupling andsignal sampling because they are not particularly high Q. Air capacitorsare fixed capacitors in which air is the dielectric material between thecapacitor's plates. It is understood that the present invention is notlimited to any particular type or number of capacitors.

As an alternative means, interdigital capacitors may be used as samplingcapacitors 204 and 203. These are a planar version of the multilayercapacitor. These capacitors have medium Q, are accurate, and aretypically less than 1 pF and may be tuned by cutting off fingers.Because interdigital capacitors have a distributed transmission linestructure, they will show multiple resonances as frequency increases.The first resonance occurs when the structure is a quarter wavelength.The Q of this structure is limited by the current crowding at the thinedges of the fingers.

As a further alternative means, metal-insulator-metal (MIM) capacitormay be used as sampling capacitors 204 and 203. A MIM capacitor, whichhas a thin insulator layer between two metal electrodes and generallythis capacitor is fabricated in semiconductor process, and thisinsulator layer provides high capacitance. Two extreme behaviors of acapacitor are that it will act as an open circuit to low frequencies orDC (zero frequency), and as a short frequency at a sufficiently highfrequency (how high is determined by the capacitor value).

The signal sampling means used in FIG. 4 may be accomplished bymeasuring the voltage from the sampling capacitors 204 and 203. The lowimpedance capacitor sampling 5 elements 201, 202 provide the stepvoltage source. The voltages are the RMS values of the “vector sum ofthe incident and reflected waves. As the source voltage varies, theinstantaneous value of voltage between the lines 106, 107 travels downthe lines 206 and 205. The ratio of the traveling voltage wave 104 tothe traveling current wave 103 is the characteristic impedance of thetransmission line. If the terminating impedance is equal to the linecharacteristic impedance, there is no wave reflected back toward thegenerator; however, if the termination resistance is any value otherthan Zo there is a reflected wave. If RL is a real impedance and greaterthan Zo, the reflected wave is 180° out of phase with the incident wave.If RL is a real impedance and is less than Zo, the reflected wave is inphase with the incident wave. The amount of variation of RL from Zodetermines the magnitude of the reflected wave. If the termination iscomplex, the phase of the reflected wave is neither zero nor 180°.Sampling the voltage at any point along the transmission line will yieldthe vector sum of the incident and reflected waves.

The sampling process is the act of turning a time continuous signal intoa signal that is time discrete or time-discontinuous. In order tomaintain frequency components of interest in the time-discontinuoussignal, the Nyquist sampling criterion is satisfied. This criterionstates that the rate of sampling of the time-continuous signal has to beat least twice as great as the frequency of the signal component withthe highest frequency which is of interest in the time-continuoussignal. A control loop can uniform supply the sampling of a continuoussignal at a constant sampling frequency.

According to an embodiment of the present invention, FIG. 5 depicts adynamic impedance matching network system 300 that is composed of a “T”network as the means for the matching network 10. The representationincludes the matching network elements 10 a, 10 b and 10 c. A “ladder”network may be use as means for the matching network 10; this is theconnection of coils and contacts used in a control circuit shown in FIG.7 one line after another that resembles. The “ladder” type circuit inFIG. 7 may comprise, but is not limited to, four inductors in series301, 302, 303, and 304. It further may include three grounded matchingcapacitors in parallel 305, 306 and 307 connected to said inductors. Thematching capacitors may be of BST type capacitors (as described in FIG.1). These lumped components are ideally suited for integrated impedancematching at low GHz frequencies. This type of matching network isfurther preferred because it may be composed of passive components suchas inductors and metal-BST dielectric-metal capacitors with high qualityfactors not generating noise or loss as with resistive networks. Thesignal collecting means used in FIG. 5 are directional couplers 101 and102. These are passive 3 or 4 port devices used to sample a portion ofthe forward (incident) signal or the reverse (reflected) signal, or both(dual directional coupler) in a RF, microwave circuit.

The purpose of the coupler 102 is to sample the incoming (mismatched)signals in one direction by means of line 110 and on reverse directionin line 109. In order to maximize coupling efficiency and to increasecoupling field, two coupling ports 109 and 110 are used. There is atransfer of RF energy from signal 104 to line 106 to the coupling meansof ports 110 and 109 into the control and power control & logic unit (PCLU) 116 which converts the input analog signals 104 into digitalsignals, senses VSWR phase and magnitude, processes the digital signalsusing an algorithm and gives it a voltage value. These values arecompared to values coming from coupler 101 though coupling ports 105 and108 into the logic and power control & 5 logic unit (PC LU) 116. Oncecompared and matched, these values are later fed through a connectionmeans 118 into a Hi Voltage Application Specific Integrated Circuit (HVASIC) 117 by which further transfers and distributes the compensatoryvoltages to the matching network elements 10 a, 10 b and 10 c.

According to a fourth advantageous embodiment, FIG. 6 depicts a dynamicimpedance matching network system 400 that may comprise of a “T” networkas the means for the matching network 10. The representation includesthe matching network elements 10 a, 10 b and 10 c. In an embodiment ofthe present invention, the signal sampling means used in FIG. 6comprises sampling capacitors 204 and 203 and may use as low impedancecapacitor sampling elements 201, 202. These are passive devices used tosample the voltage drop of a portion of the forward (incident) signal orthe reverse (reflected) signal, or both in a RF, microwave circuit atany two points in the transmission line. The sampling capacitors 204 and203 are used to estimate coupling between two circuit points to makesure a minimum of coupling is obtained.

In an embodiment of the present invention, the dynamic impedancematching network systems 100, 200, 300 and 400, may be singlestand-alone components (modules), that may be placed in the transmitchain of mobile phone radio FIG. 11. The dynamic impedance matchingnetwork systems 100, 200, 300 and 400 are advantageously self-contained,requiring only a fixed DC bias 3V from the mobile phone. The mainpurpose of the dynamic impedance matching network system 300 is tomonitor the mismatch at port 104 and dynamically change the RF match ortransfer function to maximize power transfer and minimize power lost dueto reflection into a mismatch anywhere in the RF electronic device.

The self contained dynamic impedance matching network systems ofembodiments of the present invention may be mounted onto a low-costthermally conductive dielectric substrate such as Alumina (AI203) withthe said components surface mounted on one side, then over-molded as aMulti-Chip-Module (MCM) of a typical size of 10-15 mm2 size, which isitself surface mountable on to the printed wiring board (PWB) of the RFdevise. It is understood that the present invention is not limited toany particular dielectric substrates. The dynamic impedance matchingnetwork systems may further comprise at least one DC input port 115, atleast one RF output port 109 and at least one RF input port 125(multiple ports may be used for each band). Generally, GaAs MESFETprocesses are utilized; on semi-insulating substrate and thickmetallization layers, although the present invention is not limited inthis respect. This process allows passive matching components such asspiral inductors and variable matching capacitors 313 andmetal-insulator-metal (MIM) 305, 306 and 307 capacitors with highquality factors. Fully integrated matching network elements 10 a, 10 band 10 c using spiral inductors and variable matching capacitors 313 maybe done on-chip in order to improve the reproducibility and save boardspace, although it is not required to be. These lumped passivecomponents may be ideally suited for integrated impedance matching atlow GHz frequencies.

The self-contained dynamic impedance matching network systems 100, 200,300 and 400, may be advantageously used between any two units orcomponents in any system with a varying impedance match issue. In anembodiment of FIG. 11 is a band Radio Frequency (RF) system that depictsthe multiple locations 401, 401, 403, 404 and 407 between the elementsin the system. The self-contained dynamic impedance matching networksystem may be advantageously positioned at 401 and 402 between a PowerAmplifier (PA) unit 410 and a frequency filter unit 420. The externaldigital signal processor of the mobile phone may also be used as analternative to the power control & logic unit (PC LU) 116 and the HiVoltage Application Specific Integrated Circuit (HV ASIC) 117. The PA'sin the 410 unit, the filters in the 420 unit, switches in the 430 unitand antennas 440, all may have variations in performance and impedance(unit-to-unit variations) are matched once the dynamic impedancematching network system is in line. The dynamic impedance matchingnetwork system may also be advantageously placed at 403 and 404 inbetween a frequency filter unit 420 and a switch unit 430. The dynamicimpedance matching network system may also be advantageously placed at407 between a switch unit 430 and an antenna 440. The dynamic impedancematching network system may also be advantageously placed at any of theinputs and outputs of the switch unit 430 a for example and not by wayof limitation: TxLo 403, TxHi 404, RxLI 408, Rx L2 409, TxHI 405 andTxH2 406.

Change in optimum impedance is required when operating under differentprotocols for multi-mode phones. Using this type of self-containeddynamic impedance matching network system there are no specific signalsneeded from the baseband or receiver, making it possible to be universalin nature, working in any RF devise regardless of protocol. This sameself-contained dynamic impedance matching network system works equallywell in a GSM, EDGE, CDMA or WCDMA phone, or even multi-mode phone. Thedynamic impedance matching network system will be at least capable of asingle band operation, this includes but it is not limited to; 800, 900,1800, 1900 MHz bands and the 2.1 GHz band.

The PA's in the 410 unit may be made of Gallium Arsenide(GaAs)/Heterojunction Bipolar Transistor (HBT), although the presentinvention is not limited in this respect. In a multiple band module suchas in FIG. 11, in order to compensate to optimum impedance at differentpower levels (whether the PA's in the 410 unit are saturated or linearin operation), the preferred position of the dynamic impedance matchingnetwork system may be at the final stage of the unit at positions 401and 402. The performances in each of the 2 bands (Hi and Low) are“stretched” to cover the 2 sub-bands; the self-contained dynamicimpedance matching network system compensated the PA's output andreduced the VSWR from a 10:1 to 3:1.

When positioned at 407 in FIG. 11 in a GSM band mobile phone, theresults are shown in the table of FIG. 12. The TABLE of FIG. 12 providesa graphical representation of data collected from measurements of a RFcellular phone placed on a first test environment. The test environmentconsisted of the phone placed in the pant pocket of a human beingsitting on a metal chair. Both return loss and efficiency was comparedto an OEM matching network. The data was plotted in the abscissa of thex plane the frequency in MHz vs. the y axis the antenna efficiency indB. The self-contained dynamic impedance matching network system 100 wastuned to optimize return loss at 900 MHz. Curve 530 depicts the factorymatching network and curve 520 depicts when the disclosed inventionreplaces the factory matching network. The arrow 510 depicts animprovement of >6 dB in efficiency, which means a 2×-3× gained byincorporating the disclosed invention into the phone circuit.

The TABLE of FIG. 13 is a graphical representation of data collectedfrom measurements of a RF cellular phone placed on a second testenvironment. The test environment consisted of the phone placed faced upon a metal surface with the flip cover open. Both return loss andefficiency was compared to the OEM matching network. The data wasplotted in the abscissa of the x plane the frequency in MHz vs. the yaxis the antenna efficiency in dB. The self-contained dynamic impedancematching network system 100 was tuned to optimize return loss at 900MHz. Curve 630 depicts the factory matching network and curve 620depicts when the disclosed invention replaces the factory matchingnetwork. The arrow 610 depicts an improvement of >6 dB in efficiency,which means a 2×-3× gained by incorporating the disclosed invention intothe phone circuit.

The experiments showed a significant improvement by maximizing powertransfer and reducing mismatch losses caused by component impedancevariations. Impedance variations of all components in the transmitterchain caused by temperature variations, aging or radiation hot spotscaused by VSWR problems within the phone, resulting in radiation leakingaround the display and into the user's body or hand. The experimentsalso showed that the closer the dynamic impedance matching networksystem 100 was to the antenna 440, the larger the effect and the greaterthe ability to control the impedance match. Furthermore, a significantdecrease in specific absorption rate (SAR) was seen, improving thetransmitter chain match reduced much of the radiation that was measuredduring SAR testing.

While the present invention has been described in terms of what are atpresent believed to be its preferred embodiments, those skilled in theart will recognize that various modifications to the discloseembodiments can be made without departing from the scope of theinvention as defined by the following claims.

What is claimed is:
 1. A variable circuit, comprising: a matchingnetwork coupable to a tunable device; wherein the matching networkcomprises a first port and a second port; wherein the tunable device iscoupled to one of the first port or the second port; wherein thematching network comprises one or more variable capacitors; wherein theone or more variable capacitors are operable to receive a first of oneor more variable voltage signals to cause the one or more variablecapacitors to change a first reactance of the matching network; whereinthe tunable device is operable to receive a second of one or morevariable voltage signals to cause a change in a second reactance of thetunable device; and wherein at least one of the one or more variablecapacitors comprises: a first conductor coupled to one of the first portor the second port; a second conductor; and a tunable materialpositioned between the first conductor and the second conductor, whereinat least one of the first conductor or the second conductor, or both areadapted to receive the first of the one or more variable voltage signalsto cause the change in the first reactance of the matching network. 2.The variable circuit of claim 1, wherein the change in the firstreactance of the matching network causes an increase in powertransferred from the first port to the second port or from the secondport to the first port.
 3. The variable circuit of claim 1, wherein thechange in the second reactance of the tunable device causes an increasein power transferred from the first port to the second port or from thesecond port to the first port.
 4. The variable circuit of claim 1,wherein the tunable material comprises a composition of barium strontiumtitanate.
 5. The variable circuit of claim 1, wherein the tunable devicecomprises a second tunable material.
 6. The variable circuit of claim 5,wherein the second of the one or more variable voltages is coupled tothe second tunable material to cause the change in the second reactanceof the tunable device.
 7. The variable circuit of claim 1, wherein thetunable device is a tunable antenna.
 8. A tunable circuit, comprising: amatching network coupable to an antenna, wherein the antenna comprisesan antenna member that is tunable; wherein the matching networkcomprises a first port and a second port; wherein the antenna is coupledto one of the first port or the second port; wherein the matchingnetwork comprises one or more variable reactive components; wherein theone or more variable reactive components are operable to receive a firstof one or more variable voltage signals to cause the one or morevariable reactive components to change a first reactance of the matchingnetwork; wherein the antenna member is operable to receive a second ofone or more variable voltage signals to cause the member to change asecond reactance of the antenna; wherein at least one of the one or morevariable reactive components comprises: a first conductor coupled to oneof the first port or the second port; a second conductor; and a tunablematerial positioned between the first conductor and the secondconductor, wherein at least one of the first conductor or the secondconductor, or both are adapted to receive the first of the one or morevariable voltage signals to cause the change in the first reactance ofthe matching network.
 9. The tunable circuit of claim 8, wherein atleast one of the change in the first reactance of the matching network,or the change in the second reactance of the antenna, or both causes achange in power transferred from the first port to the second port orfrom the second port to the first port.
 10. The tunable circuit of claim8, wherein each of the one or more variable reactive componentscomprises one or more variable capacitors.
 11. The tunable circuit ofclaim 8, wherein the tunable material comprises a composition of bariumstrontium titanate.
 12. The tunable circuit of claim 8, wherein theantenna member comprises a second tunable material.
 13. The tunablecircuit of claim 12, wherein the second of the one or more variablevoltages is coupled to the second tunable material to cause the changein the second reactance of the antenna.
 14. The tunable circuit of claim8, wherein at least one of the one or more variable reactive componentsof the matching network comprises the tunable material.
 15. The tunablecircuit of claim 14, wherein the tunable material comprises acomposition of barium strontium titanate.
 16. A circuit, comprising: anantenna comprising a tunable component; wherein the tunable component isoperable to receive a variable signal to cause the tunable component tochange a reactance of the antenna; and wherein the tunable componentcomprises: a first conductor coupled to the antenna; a second conductor;and a tunable material positioned between the first conductor and thesecond conductor, wherein at least one of the first conductor or thesecond conductor, or both are adapted to receive the variable signal tocause the change in the reactance of the antenna.
 17. The circuit ofclaim 16, wherein the change in the reactance of the antenna causes achange in power transferred by the antenna.
 18. The circuit of claim 16,wherein the tunable material comprises a tunable dielectric material.19. The circuit of claim 18, wherein the tunable dielectric materialcomprises a composition of barium strontium titanate.
 20. The circuit ofclaim 16, wherein the tunable material comprises a composition of bariumstrontium titanate.